This invention relates to an induction motor drive apparatus and, more particularly, to an induction motor drive apparatus which produces three-phase current commands through digital processing executed by use of a microcomputer.
Proposed methods of controlling induction motors include a so-called vector control method and field acceleration control method. Using such methods of control, the primary current of an induction motor can be controlled in terms of the instantaneous magnitude thereof to enable fine control having comparatively good response.
FIG. 1 shows an equivalent circuit of an induction motor and is useful for describing vector control. In FIG. 1, l.sub.m denotes excitation reactance, r.sub.2 represents equivalent resistance, and s designates slip. If we consider the equivalent circuit of the induction motor in this fashion, the generated torque T will be expressed by: ##EQU1## Note that .omega..sub.s represents the slip frequency. If we assume that I.sub.2 is proportional to s.multidot..omega..sub.s, then the torque T will be proportional to the secondary current and will have a torque generating mechanism similar to that of a DC motor. The following will hold from FIG. 1: ##EQU2## Accordingly, to make I.sub.2 and s.multidot..omega..sub.s proportional to each other, the excitation current I.sub.o must be made a constant quantity.
Therefore, according to vector control, the excitation current I.sub.o is held constant and only the secondary current I.sub.2 is varied, in proportion to the load torque, while perpendicularity between the excitation current I.sub.o and secondary current I.sub.2 is maintained. Also, since an error (speed error) ER between a commanded speed and the actual speed may be regarded as a torque command, the primary current I.sub.1 is defined so as to satisfy the following, in accordance with the speed error ER: EQU I.sub.1 =I.sub.o +j.multidot.k.multidot.ER (3)
In the prior art, the primary current is generated in accordance with the following method. Specifically, the method includes generating mutually perpendicular, two-phase primary current commands i.alpha. and i.beta. in accordance with the speed error ER and actual speed n, and converting these two-phase primary current commands i.alpha. and i.beta. into three-phase commands through the use of a two-phase to three-phase converter, whereby three-phase primary current commands i.sub.u, i.sub.v and i.sub.w are generated. FIG. 2 is a block diagram of an example of the prior art, in which numeral 10 denotes an induction motor, 11 a two-phase primary current generator for generating current commands i.alpha. and i.beta. of an amplitude and frequency which conform to the speed error ER and actual speed n, and 12 a two-phase to three-phase converter. As illustrated in FIG. 3, the two-phase to three-phase converter 12 is composed of resistors R.sub.1 through R.sub.4 each having a resistance of 20 K.OMEGA.., a resistor R.sub.5 of 11.55 K.OMEGA., a resistor R6 of 10 K.OMEGA., and operational amplifiers OPA.sub.1 and OPA.sub.2. In accordance with the resistance values set in this fashion, the two-phase to three-phase converter 12 performs the following vector operations to convert the mutually perpendicular two-phase currents into three-phase currents: ##EQU3##
Returning to FIG. 2, numerals 13 and 14 denote current transformers for sensing phase currents i.sub.ua and i.sub.va which flow in the U and V phases of the three-phase induction motor 10, respectively, and numeral 15 denotes an arithmetic circuit which performs the following addition operation for producing a phase current i.sub.wa to flow in the W phase of the induction motor 10: EQU -(i.sub.ua +i.sub.va).fwdarw.i.sub.wa
Numerals 16U, 16V and 16W designate current control circuits, corresponding to each of the three phases, for computing and then amplifying current differences (i.sub.u -i.sub.ua), (i.sub.v -i.sub.va) and (i.sub.w -i.sub.wa). Numeral 17 denotes a drive unit which includes a pulse-width modulator and an inverter comprising transistors.
As illustrated in FIG. 4, the pulse-width modulator PWM in the drive unit 17 includes comparators COM.sub.u, COM.sub.v, COM.sub.w, NOT gates NOT.sub.1 through NOT.sub.3, and drivers DV.sub.1 through DV6, and the inverter INV includes six power transistors Q.sub.1 through Q.sub.6 and six diodes D.sub.1 through D.sub.6. The drive unit 17 also includes a three-phase full-wave rectifier FRF. The comparators COM.sub.u, COM.sub.v and COM.sub.w compare a sawtooth signal STS, generated by a sawtooth waveform generating circuit which is not shown, with the amplitudes of three-phase alternating current signals i.sub.u ', i.sub.v ' and i.sub.w ', respectively, and produce and output with a logical value of "1" when the magnitude of STS is exceeded by i.sub.u ', i.sub.v ' or i.sub.w ', and a logic value of "0" when the magnitude of STS is greater. Thus, the comparators produce the pulse-width modulated three-phase current commands i.sub.uc, i.sub.vc and i.sub.wc dependent upon the amplitudes of i.sub.u ', i.sub.v ' and i.sub.w '. The NOT gates NOT.sub.1 through NOT.sub.3 and drivers DV.sub.1 through DV.sub.6 convert these current commands i.sub.uc, i.sub.vc and i.sub.wc into drive signals SQ.sub.1 through SQ.sub.6 to control the on/off action of each of the power transistors Q.sub.1 through Q.sub.6 comprising the inverter INV.
The foregoing conventional method has the following defects:
(1) a computation error is generated in effecting the conversion from the orthogonal two-phase system to the three-phase system;
(2) due to the computation error, the relation i.sub.u +i.sub.v +i.sub.w =0 does not hold, giving rise to a periodic undulation dependent upon motor rotation; and
(3) a function generator for generating primary currents in two phases [sin .theta. and sin (.theta.+.pi.)] and a two-phase to three-phase converter, etc., are required, thereby complicating the circuitry and raising the cost.
Another method of generating primary currents in three phases is to generate the following digitally in accordance with the speed error ER and actual speed n: EQU sin (.omega..sub.n t+.omega..sub.s t+.phi.) (7) EQU sin (.sup..omega. nt+.sup..omega. st+.phi.+2.pi./3) (8) EQU sin (.sup..omega. nt+.sup..omega. st+.phi.+4.pi./3) (9)
According to this method, a .theta.-sin .theta. characteristic, .theta.-sin (.theta.+2.pi./3) characteristic and .theta.-sin (.theta.+4.pi./3) characteristic are stored beforehand in the form of a table in a memory 22 located internally of a processor 21, illustrated in FIG. 5. The processor computes the values of .theta.=.omega..sub.n .multidot.t+.omega..sub.s .multidot.t+.phi. (where .omega..sub.n represents angular frequency, which is proportional to the rotational speed of the induction motor, .omega..sub.s represents slip frequency, and .phi. represents phase difference), sin .theta., sin (.theta.+2.pi./3) and sin (.theta.+4.pi./3). These values are applied as input signals to multiplying-type DA converters (sold under the nomenclature "DAC 08", manufactured by National Semiconductor Corporation) 23, 24 and 25. Also stored previously in the memory 22 is a relation (S-I characteristic) between the slip s and the primary current amplitude. A digital primary current I, dependent upon the slip s, is generated based on this relation and is applied to a DA converter 26 as an input signal. The output of the DA converter 26 is applied as an input to each of the multiplying-type DA converters 23, 24 and 25 which then produce the following as respective output signals: EQU i.sub.u =I.multidot.sin .theta., i.sub.v =I.multidot.sin (.theta.+2.pi./3), i.sub.w =I.multidot.sin (.theta.+4.pi./3)
In FIG. 5, portions similar to those shown in FIG. 2 are designated by like reference characters and a detailed description thereof will not be provided.
The above mentioned, however, possesses the following defects:
(1) the relation i.sub.u +i.sub.v +i.sub.w =0 is not produced with accuracy, resulting in a periodic undulation, and
(2) tables and DA converters are required for all three phases, thereby raising cost.